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? 2004 microchip technology inc. ds00912a-page 1 AN912 introduction this application note builds on application note an232 low frequency magnetic transmitter design (ds00232). it covers the design process to implement lf talkback functionality. an232 covers some of the magnetism basics and design principles to implement the drive circuitry. lf talkback generally refers to the process in which a transponder can communicate back to a magnetic transmitter base station by loading the generated magnetic field. by measuring the small changes in the transmitter coil's voltage, used to gener- ate the field, the communications? data is extracted. lf talkback is commonly used in rfid, automotive transponders, active transponders, and many other bidirectional lf communications topologies. this document will cover the different stages needed to implement a typical lf talkback system and explain the process in choosing the different stage characteris- tics. it explains the various performance and cost trade- offs made for the reference design and how it can be adapted to better suit the readers needs. main building blocks figure 1 shows the main building blocks that make up the lf talkback system described in this document. the base station generates a strong magnetic field by setting up resonance in a serial resonant tank. the circulating energy in the resonant tank typically gener- ates 300v peak-to-peak voltage across the transmitting antenna coil at 125 khz. the transponder, whether active or passive, is magnetically coupled to the base station?s transmitting coil and the transponder?s magnetic loading has a small effect on the quality factor (q) of the transmitter resonant tank. talkback is accomplished by changing or modulating the magnetic loading and can be observed as small voltage changes across the base station's resonant transmitter coil. the difficulty is to detect a few mv of modulation on the 300v peak-to-peak carrier. figure 1: author: ruan lourens microchip technology inc. v ref data transponder base station m peak detector dc decouple low pass filter data slicer data designing lf talkback for a magnetic base station
AN912 ds00912a-page 2 ? 2004 microchip technology inc. a high voltage peak detector is used to extract the basic envelope of the base station's resonant tank. the output of the peak detector will be 150 v dc with about 2v peak-to-peak of carrier ripple at 125 khz and then about 2 mv of modulated signal. the modulation signal strength is mostly dependent on the distance between the transponder and the transmitter coil as the magnetic coupling decreases to the third power of the distance between the two devices. the next stage is a passive high-pass filter to decouple or block the high dc voltage. the dc extracted voltage is then fed into a low-pass filter, leaving the required modulating signal. the last stage is the data slicer that compares the modulating signal to some reference to extract the original signal sent by the transponder. lf talkback receiver can be thought of as detecting and decoding an amplitude modulation (am) signal that has a very low modulation index on a relatively large carrier. system assumptions the lf talkback system designed in this document is targeted for a lf base station that has the following characteristics and is based on the design as per an232: the lf talkback signal is amplitude modulated at 200 s multiples. this is also referred to as the basic pulse element period or t e . the tank is driven by a 12v half-bridge driver. the tank inductance is 162 h and the resonant capacitor is 10 nf with a resonant frequency of 125 khz. the tank q is 25. as a result, the tank or carrier voltage is 300v peak-to-peak or 150v 0-to-peak. transponder induced modulation of 2 mv in magnitude needs to be detected. to get an understanding of the impedances involved, lets consider the following: using equation 1, the equivalent parallel resistance of the tank is 3.18 k ? . the additional parallel impedance that a transponder represents to induce a 2 mv signal on the tank is in the order of 500 m ? . what the lf talkback system detects is the result of a 500 m ? resistor being switched in and out in parallel with the tank at the data rate. therefore, it is very important that the peak detector have a high- impedance at the data rate to maintain good sensitivity. equation 1: the effective parallel impedance of a resonant tank the peak detector there are a number of aspects to consider in designing a peak detector for this application: 1. the peak detector has to be able to operate at the high voltages of the resonant tank. 2. maintain a good tank q or, in other words, it should not add unnecessary loading on the main resonant tank. if it does load the tank, it will result in a lower modulation voltage induced by the transponder. 3. reduce carrier ripple as far as possible. 4. maintain the modulation signal. 5. have a fast large swing dynamic response and be able to settle quickly after the field is turned on. 6. cost of the system. some of the peak detector requirements are conflicting and as a result, the designer has to find an acceptable compromise with the final system performance in mind. one can sacrifice a specific parameter and make up for it in a later stage where optimization of that aspect is easily accomplished. as an example to optimize requirement 3, one needs to increase the size of the capacitor c2 (figure 2), but that will negatively affect requirements 2, 4 and 5 if a passive peak detector is used. an active peak detector could have solved the conflict, but at the 600v swing, one has little choice but to use a passive peak detector while maintaining a low-cost design. a relatively low capacitance value is chosen for c1 of 1 nf. this maintains the dynamic response requirement for settling quickly after the field is applied and does not load the tank unnecessarily. capacitor c2 should have at least a 300 v dc peak rating and a high tolerance capacity is acceptable to save cost. an ultra fast diode is required in the peak detector with a 400v or better rating and low junction capacitance. a uf1005 diode was chosen, it has a 600v rating and 10 pf of junction capacitance. figure 2: r parallel = 2 lf c q l = tank inductance in h = 162 h fc = center frequency of tank = 125 khz q = tank quality factor = 25 l1 c1 c2 r1 d1 hv-env
? 2004 microchip technology inc. ds00912a-page 3 AN912 the envelope detector with only d1 and c2 has a greatly different response to increasing and decreasing voltage amplitudes of the resonant tank. the voltage designated by the signal hv_env (figure 2) rises quickly with increasing tank amplitudes because d1 has a low-impedance in forward conduction. the tank voltage decreases slower when the tank amplitude is lowered because c2 can only discharge through d1, which has a high-impedance in the reverse direction. the situation can be remedied to some extent by the introduction of r1 which helps to discharge c2, but the value of r1 should be high enough to maintain a good tank q as per requirement 2 above. a 10 m ? value for r1 works well, but note that r1 needs to be implemented as a series of two resistors. this is done to stay within the safe voltage range of 0805 resistors are used. the 125 khz carrier ripple voltage, without r1, is about 2v peak-to-peak and is due to the junction capacitance and reverse leakage of d1. the addition of r1 has little effect on the ripple voltage, but does improve the detectors dynamic performance at the data rate. the carrier ripple voltage will be filtered out at a later stage where a more effective solution can be implemented. the dc decoupler conflicts the hv_env signal (figure 2) consists of three main components: 1. a 150v dc signal, as a result of the peak detector. 2. 2v peak-to-peak ripple voltage at the carrier frequency. 3. the modulated data signal at a t e of 200 s and a 2 mv peak-to-peak amplitude, highest funda- mental harmonic content is at 2.5 khz [1/(2*200 us)], irrespective of the modulation scheme used (i.e., manchester, pwm etc.). the aim of the decoupling stage is to reject the high dc voltage without adding unnecessary loading to the tank via the peak detector. it should also have a fast dynamic response and stabilize quickly after the tank is ener- gized. the dynamic response of the lf talkback system is the major design hurdle to overcome as far as the decoupling stage is concerned. the problem is aggra- vated when the transponder needs to communicate on the lf link soon after the base station communicated with the transponder. the base station typically uses on off keying (ook) modulation to communicate to the transponder. this means the tank resonance is completely halted and then started up to transfer data via the magnetic link. the decoupling stage experiences large ?step? responses as data is transmitted to the transponder. the tank can ramp up to its full resonant amplitude in 100 s to 400 s depending on the drive system used. figure 3: the system can be simplified as shown in figure 3. the output of the peak detector can be simplified as the step response source with a 150v amplitude that also has the carrier and data signals superimposed on it as described earlier. the output response of the decoupling stage is given by equation 2. this is also the input signal to the low-pass filter. equation 2: it is useful to think in terms of (rc time constant) because the voltage across the resistor reduces by a factor of 0.368 as every second elapses. the exponential decay curve, for the voltage across r, is shown in figure 4 and indicates that the initial voltage decays rapidly, but settles out slower as the voltage is reduced across the resistor. the system must be allowed to settle for a long enough period so that the step response voltage has reduced to a voltage that is smaller than the modulation voltage. the required value for rc, or , can be calculated using equation 3, based on the following assumptions: the system needs to be able to start lf communi- cations 200 s after the resonant tank has stabilized. the decoupler should settle to at least half the data modulation voltage. hv-env c r lp filter v = 150 e -t/ = rc
AN912 ds00912a-page 4 ? 2004 microchip technology inc. figure 4: equation 3: using equation 3, was calculated to be 16.78 s. the question now is how will the data signal be affected by the decoupling stage? the decoupling stage, shown in figure 3, is also a high-pass filter and it was calculated that the rc time constant needs to be 16.78 s to satisfy the transient response requirement. the 3 db cutoff frequency, for a , of 16.78 s is calculated as 9.48 khz using equation 4. this means that the decoupling stage will only pass one quarter of the original data signal at 2.5 khz, which is not desirable from a signal-to-noise ratio perspective. equation 4: from a data signal conservation, or high-pass filter point-of-view, should be at least 64 s. the conflicting requirement shows that a basic high-pass filter is not sufficient as a decoupler unless either dynamic response or data signal strength is sacrificed. 160 140 120 100 80 60 40 20 0 01 2 34 5 voltage ln( vo/v settle ) t settle t settle = 200 s v o = 150 v v settle = 1 mv = f c = 1 2
? 2004 microchip technology inc. ds00912a-page 5 AN912 an improved decoupler from the previous section, it is clear that a high-pass filter is needed with either a controllable or a nonlinear that is based on the voltage across the output of the decoupler. both approaches will be covered and the latter solution is shown in figure 5. figure 5: the addition of the two diodes, shown in figure 5, results in a nonlinear with respect to voltage because it effectively lowers the r component of whenever the voltage is either above 3.1v or below -3.1v. in a practi- cal circuit, the diodes will start conducting when the tank is turned on and the voltage, across the resistor r, is around 3 volts, after the tank has stabilized. previously, was calculated with an initial voltage of 150 v dc , but if the calculation is repeated with an initial voltage of 3 v dc , then the required comes to 25 s. the rc time constant is improved by a significant factor from 16.78 s to 25 s with the additional diodes, however, it is still not in the 64 s ball park. the diodes have the additional advantage in that they protect the low-pass filter from the large positive and negative voltages that develop across the resistor during tank transient periods. the final part to solving the time constant problem is to add an additional resistor via a switch, as shown in figure 6. the switch is closed to reduce from 64 s to 25 s during transient periods and opened while data is received via the lf talkback link. figure 6: the final part of the decoupling stage is to lower the output impedance by adding an active buffer in the form of an inverting amplifier that has an input resis- tance equal to r1. the use of an inverting amplifier has the additional advantages that it can add gain and a single order low-pass filter to the decoupler, as shown in figure 7. the gain is equal to the ratio of r3/r1, and the low pass cutoff frequency is set by r3 and c2, as per equation 4. the low pass cutoff frequency should be chosen at least two decades above the main low- pass filter, otherwise it will have an undesirable effect on the envelope response. for single ended 5v designs, the gain should be limited to about 10 db to avoid amplifier saturation due to carrier ripple and data modulation. figure 7: hv-env c r lp filter 2.5v -2.5v hv-env c r1 lp filter 2.5v -2.5v r2 s1 hv-env r2 s1 - + outpu t c2 r1 r3 c1
AN912 ds00912a-page 6 ? 2004 microchip technology inc. the low pass filter stage the output signal from the decoupling stage consists of the 125 khz carrier ripple and the modulated data signal, if one ignores the dynamic response signal. the carrier ripple is about 300 mv peak-to-peak. the data is 4 mv peak-to-peak with 6 db of gain of the decoupler and a cutoff frequency at about 10 khz. the aim of the low-pass filter stage is to amplify the data signal at 2.5 khz and to filter out the carrier ripple in the most effective manner. the three most common active filter topologies used are the chebyshev, butterworth and bessel filters. the chebyshev filter has the steepest transition from pass band to stop band, but has ripple in the pass band. the butterworth filters have the flattest pass band response, but does not have such a steep transition as the chebyshev. the bessel filter has a linear phase response with a smooth transition from pass to stop band. it seems the chebyshev filter would best be suited for this application, but the frequency response does not tell the whole story. the data signal is amplitude modulated and the tank has steep transient response dynamics. as a result, the filter should have a stable and flat transient response. the chebyshev filter has a very sharp frequency cutoff response, but has the worst transient response of the three filter topologies. the chebyshev filter also has an underdamped step response with overshoot and ringing. the butterworth filter has a better transient response, but still some overshoot. the bessel filter has the worst response from a frequency perspective, but has the best transient response as a result of its linear phase characteristics. there are of course other active filter topologies such as elliptical, state variable, biquad and more, but a bessel filter has adequate performance for the application. the data signal, in this example, has maximum modulation frequency of 2.5 khz or a t e of 200 s. a bessel filter, with a cutoff frequency of 1/(2.2t e ) = 2.27 khz, would be ideal from a noise rejection point of view, but a 2.5 khz cutoff was chosen to minimize sym- bol overlap. the target is to design a filter with sufficient performance using a single operational amplifier in order to reduce the system cost. a dual operational amplifier can then be used because the decoupling stage also uses an amplifier. a third order bessel filter can now be implemented with the remaining amplifier. the filter gain is the final aspect to specifying the bessel filter. using microchip's filterlab ? program, one can get the response for a unity gain ? 2.5 khz, 3d order bessel filter. at 125 khz, the filter has 93 db of attenuation and the input ripple amplitude is 300 mv peak-to-peak. assuming the filter should have an output ripple of no more then 1 mv peak-to-peak with 12 db of headroom for noise, coupled through the supply line, then one needs at least 62 db of attenua- tion. this leaves 31 db of allowable gain from the third order filter. for the design, a gain of 20 or 26 db was chosen, leaving some additional headroom for ripple rejection. the 3d order low-pass bessel filter is shown in figure 8 and has a fc = 2.5 khz and 26 db of gain. please note that the circuit shown in figure 8 has a fairly high output impedance at the data rate, but the output of the filter will be driving a high-impedance load, and this is therefore acceptable. figure 8: input - + 4.87k 10 nf outpu t 78.7k 16.5k 150 pf 10 nf 3.92k
? 2004 microchip technology inc. ds00912a-page 7 AN912 data slicer the data slicer is essentially a comparator with some input hysteresis voltage to reduce the influence of noise. the overall system gain of the decoupler and the low-pass filter, at 2.5 khz, is about 29 db or a factor of 28, and the system should be able to detect the 2 mv data signal. the headroom between the hysteresis and data signal was chosen to be about 9 db or a factor of 2.8. this means that the minimum input voltage to overcome the data slicer hysteresis is about 700 v. this translates to 20 mv of hysteresis for the data slicer. most comparators have some deliberate hyster- esis to improve noise stability and this amount should be extracted from the required hysteresis when calcu- lating the amount of required feedback. figure 9 shows a typical hysteresis circuit and equation 5 can be used to calculate the amount of hysteresis for a single-ended circuit. equation 5: figure 9: for example, if a comparator with 10 mv of offset and hysteresis is used, then an additional 10 mv of hyster- esis should be added. the resistor r2 is calculated to be 5 m ? for a v dd of 5 v dc and r1 = 10 k ? . an example system a complete circuit with layout, based on the foregoing design study, is shown in this section. the circuit diagram is shown in figure 11. the top and bottom layout for the printed circuit board is shown in figure 12. the pic16f648a was chosen for the application, it has two comparators, a usart, eeprom and 4k of flash program memory. the pic16f648a can be substituted with its smaller program memory equivalents, the pic16f627a or pic16f628a. the filter examples have been converted to operate from a single 5 v dc supply. the 2.5 v dc virtual ground is provided by the voltage divider consisting of r23 and r24, shown in figure 11. the reference voltage does not have to be actively buffered, it is lightly loaded. a 0.1 f decoupling capacitor c10 is sufficient for noise reduction. a tc4422 fet driver, u1, drives the resonant tank consisting of l1 and c2. the tank generates a strong magnetic field and the voltage at the test pin tp1 can reach 320v peak-to-peak. the main antenna, l1, is an air-cored inductor with a 25 mm radius and 41 turns of 26-gage wire, and has a 162 h inductance. the inductor l2 and capacitors c3 and c4 are not popu- lated and are added to the printed circuit board to test alternative antennas. the peak detector consists of d1, c5, r1, and r2, and is connected to the decoupling stage via c6. the rc time constant of the decoupling tank is set by c6 and r4 to 177 s, which is substan- tially longer than the minimum filter requirement of 64 s. resistor r3 is used to change the decoupler's time constant to 11 s by changing rb7 from a high- impedance input to an output. the decoupler buffer, u2:a, has a gain of 6 db and a low pass cutoff frequency at 9.8 khz, set by r5 and c8. the r22 resistor is used to ensure the proper dc bias of the stage, but does not have a significant effect on the overall sensitivity. the output of the decoupler is connected to the input of the low pass bessel filter and one of the pic16f648's comparators. the remaining op amp, u2:b, is used for the bessel filter. u2 is a dual mcp6002 op amp that has a gbwp of 1 mhz. the filter components should have better tolerances than the high voltage components and 1% resistors. the 5% np0 capacitors are recommended. v hyst = r 1 r 2 v dd input - + output r1 v ref r2
AN912 ds00912a-page 8 ? 2004 microchip technology inc. the pic16f648a has various comparator options. figure 10 shows the topology that was chosen for this application. the main filter output signal ?env_in? is connected to comparator c1 via ra0. resistor r10 was placed in series with the output of the filter to have 10 k ? impedance. together with the 4.99 m ? resistor, r11 adds an additional 10 mv of hysteresis. the comparator has a combined offset and hysteresis of 10 mv, in the worst case, making for a total of 20 mv of hysteresis, in the worst case, and about 15 mv on average. it should be noted that the output of compar- ator c1 has to be inverted by setting bit c1inv, in the cmcon register. the output inversion is needed to result in positive feedback, via r11, as is shown in figure 9. at first glance, it seems as if r10 can be removed and r11 changed to a 2.43 m ? resistor, but the capacitor c12 will cause delay and that can lead to instability. figure 10: there are additional aspects around the decoupler that need to be explained for the system as it is imple- mented. the port pin rb7 is essentially an open circuit when it is configured as an input and the input voltage is between v dd (5v) and ground. all the general purpose i/o pins have internal esd protection diodes that become conductive when a pin voltage is forced outside the v dd to ground range. this has the effect that the rc time constant for the decoupling stage is reduced to 11 s from 177 s whenever the ?bias? signal is about 0.6v above v dd , or below ground even if rb7 is configured as an input. the addition of r3 works well, but keep in mind, the stable dc voltage for signal ?a?, shown in figure 11, is 2.5 v dc and the signal ?bias? is either 5v or ground. one can implement one of two approaches to correctly bias the signal at point ?a?. the first solution is to toggle rb7 between high and low with a 50% duty cycle at 20 khz or more. this is equivalent to connecting the ?bias? signal to the desired 2.5 v dc . this is only done for a short period after the tank is turned on or off, to force the decoupler to stabilize faster than it would with just r4. the second approach is to force the signal ?a? in the required direction. the voltage at ?a? will go above v dd if the tank is turned on after it has been turned off for some time. the ?bias? signal can be grounded during the turn-on transient period until the voltage at point ?a? reaches the desired 2.5 v dc or v ref . by monitoring either of the comparator output signals, it is possible to detect when the voltage at point a goes through v ref . pin rb7 can be turned into an input as soon as the cross over is detected resulting in a decoupler rc time constant of 177 s. the filters introduce delay that cause some overshoot of the voltage at point ?a?. the overshoot can be resolved by allowing some additional stabilizing time with r4, before lf communication is interpreted as data. - + - + two common reference comparators with outputs cm2:cm0 = 110 c1 c2 v in - v in + a d c1v out c2v out v in - v in + a a open drain ra0/an0 ra3/an3/cmp1 ra1/an1 ra2/an2/v ref ra4/t0cki/cmp2
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AN912 ds00912a-page 10 ? 2004 microchip technology inc. appendix a: schematics figure 11: lf base station notes: unless otherwise specified; resistance values are in ohms. resistors are 1% tolerance. capacitance values are in uf. smt resistors are size 1206 and 1/8w. device names and numbers shown here are for reference only and may differ from the actual number. items labeled with a are unpopulated. items labeled with b are socketed and populated. high-voltage section +5v c1 0.1 uf 1 in pwm 3 v dd 6 v dd 5 out gnd gnd 2 4 u1 tc4422 l1 10-00189 l2 do5022p tp1 a coarse_env_in c2 10 nf 400v p3476-nd a c3 .200ls c4 .200ls uf1005 d1 tp2 c5 1.0 nf 500v 1412ph-nd a c6 2.2 nf 500v a r1 4.99m r2 4.99m a r3 4.99k bias tp3 r4 80.6k r22 4.99m v ref v ref 3 2 -in +in 8 4 v dd v ss out u2:a mcp6002/sn c8 100 pf r5 162k +5v c7 0.1 uf 1 tp4 r6 3.92k +5v r23 49.9k v ref r24 49.9k c9 10 nf r7 16.5k v ref c10 0.1 uf 5 6 -in +in out 7 r8 78.7k c11 150 pf u2:b mcp6002/sn r9 4.87k tp5 c12 10 nf r10 5.11k r11 4.99m env_in env_out coarse_env_out tp6 tp7
? 2004 microchip technology inc. ds00912a-page 11 AN912 figure 12: lf base station (continued) +5v 1 uf c21 c20 1 uf 16 u5 max232cpe 2 v cc tx rx r14 1k r15 1k 11 10 12 9 1 3 6 v+ t1in t2in r1out r2out c1+ c1- v- c18 1 uf c17 1 uf 15 gnd t1out t2out r1in r2in c2+ c2- 14 7 13 8 4 5 c19 1 uf j1 1 2 3 4 5 6 7 8 9 de-9s (fem) reset sw2 1 23 4 mom-no +5v r20 4.7k r21 470 mclr +5v r19 4.7k r18 470 rb0 sw1 mom-no 3 4 2 1 a j3 1 2 3 4 5 6 rbo rx tx rb5 rb6 bias c23 20 pf y1 20.0 mhz 311-1153-1-nd osc1 osc2 c22 20 pf c24 0.1 uf +5v coarse_env_in env_in osc1 osc2 bias rb5 rb6 rf_in 18 17 16 15 14 13 12 11 10 pic16f648a/p ra1 ra0 osc1 osc2 v dd rb7 rb6 rb5 rb4 ra2 ra3 ra4/to mclr v ss rbo/int rb1 rb2 rb3 1 2 3 4 5 6 7 8 9 u3 b v ref env_out coarse_env_out mclr rb0 rx tx pwm +12v j2 4p-din 1 2 3 4 5 6 7 power dynamic mdc-034 c13 560 uf 25v p11220-nd vr1 78l05 out in gnd 1 3 2 c14 47 uf 10v p11180-nd d2 grn power +5v r12 270 rf_in r13 1k wire antenna a1 6.8" 1 c16 0.1 uf c15 10 uf 6.3v +5v u4 1 2 3 7 rf_+v cc rf_gnd data_in rf_gnd 10 11 12 13 14 15 nc af_+v cc af_gnd af_+v cc tp data_out af_+v cc rr8 433.92 mhz +5v +5v r16 270 d3 red rb5 rb6 d4 grn r17 270
AN912 ds00912a-page 12 ? 2004 microchip technology inc. figure 13: bottom side figure 14: top side 05-01 xxxx rev. a bottom side 05-01 xxxx rev. a top side
? 2004 microchip technology inc. ds00912a-page 13 AN912 figure 15: top mask 05-01 xxxx rev. a top mask c21 c20 c18 c19 rs232 c17 r14 r15 u5 j1 j2 d2 r12 c13 vr1 c14 power d5 c15 c16 r13 c22 c23 c24 reset r20 r21 r19 rb0 r18 a1 d3 r16 d4 r17 u4 rb0 rb1 rb2 rb5 c9 rb6 rb7 filter tp7 tp6 tp5 r11 r10 u3 r9 c11 r8 r7 r6 tp4 tp3 r5 c8 r4 r3 j3 c12 c7 u2 r23 r24 high voltage section u1 d1 c2 c3 c4 c5 r1 c6 c10 r22 r2 d1 tp2 tp1 l1 l2
AN912 ds00912a-page 14 ? 2004 microchip technology inc. notes:
? 2004 microchip technology inc. ds00912a-page 15 information contained in this publication regarding device applications and the like is intended through suggestion only and may be superseded by updates. it is your responsibility to ensure that your application m eets with your specifications. no representation or warranty is given and no liability is assumed by microchip technol ogy incorporated with respect to the accuracy or use of such information, or infringement of patents or other intellectual property rights arising from such use or otherwise. use of mi crochip?s products as critical components in life support syst ems is not authorized except with express written approval by microchip. no licenses are conveyed, implicitly or ot herwise, under any intellectual property rights. trademarks the microchip name and logo, the microchip logo, accuron, dspic, k ee l oq , mplab, pic, picmicro, picstart, pro mate and powersmart are registered trademarks of microchip technology incorporated in the u.s.a. and other countries. amplab, filterlab, micro id , mxdev, mxlab, picmaster, seeval, smartshunt and the embedded control solutions company are registered trademarks of microchip technology incorporated in the u.s.a. application maestro, dspicdem, dspicdem.net, dspicworks, ecan, economonitor, fansense, flexrom, fuzzylab, in-circuit serial programming, icsp, icepic, microport, migratable memory, mpasm, mplib, mplink, mpsim, pickit, picdem, picdem.net, pictail, powercal, powerinfo, powermate, powertool, rflab, rfpic, select mode, smartsensor, smarttel and total endurance are trademarks of microchip technology incorporated in the u.s.a. and other countries. serialized quick turn programming (sqtp) is a service mark of microchip technology incorporated in the u.s.a. all other trademarks mentioned herein are property of their respective companies. ? 2004, microchip technology incorporated, printed in the u.s.a., all rights reserved. printed on recycled paper. note the following details of the code protection feature on microchip devices: microchip products meet the specification cont ained in their particular microchip data sheet. microchip believes that its family of products is one of the mo st secure families of its kind on the market today, when used i n the intended manner and under normal conditions. there are dishonest and possibly illegal methods used to breach the code protection feature. all of these methods, to our knowledge, require using the microchip produc ts in a manner outside the operating specif ications contained in microchip's data sheets. most likely, the person doing so is engaged in theft of intellectual property. microchip is willing to work with the customer who is concerned about the integrity of their code. neither microchip nor any other semicondu ctor manufacturer can guarantee the security of their code. code protection does not mean that we are guaranteeing the product as ?unbreakable.? code protection is constantly evolving. we at microchip are comm itted to continuously improving t he code protection features of our products. attempts to break microchip?s c ode protection feature may be a violation of the digital millennium copyright act. if such acts allow unauthorized access to your software or other copyrighted work, you may have a right to sue for relief under that act. microchip received iso/ts-16949:2002 quality system certification for its worldwide headquarters, design and wafer fabrication facilities in chandler and tempe, arizona and mountain view, california in october 2003. the company?s quality system processes and procedures are for its picmicro ? 8-bit mcus, k ee l oq ? code hopping devices, serial eeproms, microperipherals, nonvolatile memory and analog products. in addition, microchip?s quality system for the design and manufacture of development systems is iso 9001:2000 certified.
ds00912a-page 16 ? 2004 microchip technology inc. americas corporate office 2355 west chandler blvd. chandler, az 85224-6199 tel: 480-792-7200 fax: 480-792-7277 technical support: 480-792-7627 web address: http://www.microchip.com atlanta 3780 mansell road, suite 130 alpharetta, ga 30022 tel: 770-640-0034 fax: 770-640-0307 boston 2 lan drive, suite 120 westford, ma 01886 tel: 978-692-3848 fax: 978-692-3821 chicago 333 pierce road, suite 180 itasca, il 60143 tel: 630-285-0071 fax: 630-285-0075 dallas 4570 westgrove drive, suite 160 addison, tx 75001 tel: 972-818-7423 fax: 972-818-2924 detroit tri-atria office building 32255 northwestern highway, suite 190 farmington hills, mi 48334 tel: 248-538-2250 fax: 248-538-2260 kokomo 2767 s. albright road kokomo, in 46902 tel: 765-864-8360 fax: 765-864-8387 los angeles 18201 von karman, suite 1090 irvine, ca 92612 tel: 949-263-1888 fax: 949-263-1338 san jose 1300 terra bella avenue mountain view, ca 94043 tel: 650-215-1444 fax: 650-961-0286 toronto 6285 northam drive, suite 108 mississauga, ontario l4v 1x5, canada tel: 905-673-0699 fax: 905-673-6509 asia/pacific australia suite 22, 41 rawson street epping 2121, nsw australia tel: 61-2-9868-6733 fax: 61-2-9868-6755 china - beijing unit 706b wan tai bei hai bldg. no. 6 chaoyangmen bei str. beijing, 100027, china tel: 86-10-85282100 fax: 86-10-85282104 china - chengdu rm. 2401-2402, 24th floor, ming xing financial tower no. 88 tidu street chengdu 610016, china tel: 86-28-86766200 fax: 86-28-86766599 china - fuzhou unit 28f, world trade plaza no. 71 wusi road fuzhou 350001, china tel: 86-591-7503506 fax: 86-591-7503521 china - hong kong sar unit 901-6, tower 2, metroplaza 223 hing fong road kwai fong, n.t., hong kong tel: 852-2401-1200 fax: 852-2401-3431 china - shanghai room 701, bldg. b far east international plaza no. 317 xian xia road shanghai, 200051 tel: 86-21-6275-5700 fax: 86-21-6275-5060 china - shenzhen rm. 1812, 18/f, building a, united plaza no. 5022 binhe road, futian district shenzhen 518033, china tel: 86-755-82901380 fax: 86-755-8295-1393 china - shunde room 401, hongjian building, no. 2 fengxiangnan road, ronggui town, shunde district, foshan city, guangdong 528303, china tel: 86-757-28395507 fax: 86-757-28395571 china - qingdao rm. b505a, fullhope plaza, no. 12 hong kong central rd. qingdao 266071, china tel: 86-532-5027355 fax: 86-532-5027205 india divyasree chambers 1 floor, wing a (a3/a4) no. 11, o?shaugnessey road bangalore, 560 025, india tel: 91-80-2290061 fax: 91-80-2290062 japan benex s-1 6f 3-18-20, shinyokohama kohoku-ku, yokohama-shi kanagawa, 222-0033, japan tel: 81-45-471- 6166 fax: 81-45-471-6122 korea 168-1, youngbo bldg. 3 floor samsung-dong, kangnam-ku seoul, korea 135-882 tel: 82-2-554-7200 fax: 82-2-558-5932 or 82-2-558-5934 singapore 200 middle road #07-02 prime centre singapore, 188980 tel: 65-6334-8870 fax: 65-6334-8850 taiwan kaohsiung branch 30f - 1 no. 8 min chuan 2nd road kaohsiung 806, taiwan tel: 886-7-536-4818 fax: 886-7-536-4803 taiwan taiwan branch 11f-3, no. 207 tung hua north road taipei, 105, taiwan tel: 886-2-2717-7175 fax: 886-2-2545-0139 europe austria durisolstrasse 2 a-4600 wels austria tel: 43-7242-2244-399 fax: 43-7242-2244-393 denmark regus business centre lautrup hoj 1-3 ballerup dk-2750 denmark tel: 45-4420-9895 fax: 45-4420-9910 france parc d?activite du moulin de massy 43 rue du saule trapu batiment a - ler etage 91300 massy, france tel: 33-1-69-53-63-20 fax: 33-1-69-30-90-79 germany steinheilstrasse 10 d-85737 ismaning, germany tel: 49-89-627-144-0 fax: 49-89-627-144-44 italy via quasimodo, 12 20025 legnano (mi) milan, italy tel: 39-0331-742611 fax: 39-0331-466781 netherlands p. a. de biesbosch 14 nl-5152 sc drunen, netherlands tel: 31-416-690399 fax: 31-416-690340 united kingdom 505 eskdale road winnersh triangle wokingham berkshire, england rg41 5tu tel: 44-118-921-5869 fax: 44-118-921-5820 01/26/04 w orldwide s ales and s ervice

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